1. Field of the Invention
This invention relates generally to radio frequency receivers, and more specifically to reducing second order intermodulation distortion in a direct conversion receiver.
2. Related Art
A receiver uses the frequency response of a low noise amplifier (LNA), a surface acoustic wave (SAW) filter and a duplexer to attenuate signals that are far from a center frequency of the receiver sufficiently enough to not corrupt a desired signal. If the LNA and the SAW filter are removed from the analog line-up of the receiver, problems that can detrimentally affect the performance of the receiver may arise. In a transceiver, which comprises a transmitter and a receiver, one such problem is a signal transmitted by the transmitter leaking into a receive path of the receiver. In a receiver with only a duplexer to isolate the receiver from the transmitter, there is considerably less attenuation at the transmitted frequency. A receiver that lacks an LNA and a SAW filter requires additional and/or tighter constraints on at least some of the non-idealities in the analog line-up of the receiver. One example on which a tighter constraint is necessary is the second order intercept point (IP2) of the mixer. Without a sufficiently high IP2 of the mixer, the presence of second order intermodulation distortion (IMD2) substantially reduces the sensitivity of the receiver.
Most cellular wireless transceivers use a direct-conversion receiver because a high level of integration can be obtained. However, a direct-conversion receiver requires a high input-related second order intercept point (IIP2), which is the theoretical input level at which the power of the IMD2 products are equal in power to the power of a desired signal. FIG. 1 shows a simplified functional block diagram of a portion of a typical known direct-conversion, third-generation (3G) receiver 102 that lacks both an LNA and a SAW filter. The receiver 102 may be part of a transceiver 101 that includes a transmitter 103. The receiver 102 comprises a transconductance amplifier (TCA) 110 that amplifies a received signal, and a local oscillator 112. The TCA 110 is coupled to an antenna 106 via a duplexer 108. If the receiver 102 had included an LNA and a SAW filter, they would have been typically present in the analog line-up between the duplexer 108 and the TCA 110. Next, an in-phase (I-phase) mixer 114 and a quadrature-phase (Q-phase) mixer 115 are employed to convert the RF signal to a zero-IF, or baseband, signal. The mixers 114 and 115 are driven by the local oscillator 112. The frequency of the local oscillator 112 is controlled by a 3G receiver RF phase locked loop 116 that is coupled to a divide-by-two circuit 117. One output of the local oscillator 112 is phase shifted by 90° to provide an I-phase component and a Q-phase component of the received signal. The output of each mixer 114 and 115 is coupled to a biquad filter 118 and 119. The biquad filters 118 and 119 attenuate higher frequency signals. Each biquad filter 118 and 119 is coupled to a baseband amplifier 122 and 123. The output of each baseband amplifier 122 and 123 is coupled to an analog lowpass filter 124 and 125. The analog lowpass baseband filters 124 and 125 attenuate adjacent channel blockers and attenuates higher frequencies. The output of each lowpass filter 124 and 125 is coupled to a sigma-delta analog-to-digital (A/D) converter 130 and 131. The output of the A/D converters 130 and 131 provides I samples and Q samples, respectively, to decimation filters 140 and 141 for further processing by the receiver 102.
In the receiver 102, transmitted signals are attenuated via the duplexer 108 by approximately 50 dB; nevertheless, attenuated transmitted signals leak into the receive signal path prior to the TCA 110. For example, in the receiver 102, the duplexer 108 attenuates a strong transmitted signal of +25 dBm (316 milliwatts) located at 190 MHz from the center frequency by only 50 dB, thus resulting in a signal of −25 dBm (3.16 μwatts) at the input of the TCA 110. This −25 dBm signal creates strong IMD2 products that land on the desired signal, thus producing co-channel interference. Without a sufficiently high IP2, the IMD2 can greatly detrimentally affect the sensitivity of the receiver 102.
FIG. 2 shows an example of how the IP2 of the mixer 114 and 115 in the receiver 102 can vary due to one or more factors. FIG. 2 is an idealized graph showing how the IP2 of the mixer 144 and 115 can vary due to manufacturing processes and/or change in temperature. Any mismatch between differential signals I+ and I− in mixer 114 causes a reduction in the IP2 from the optimal IP2 (located at the center point of the graph of IP2). When I+ and I− are not matched, due to variations in manufacturing processes or due to temperature changes during operation of the receiver 102, or due to both causes, there is a worse IP2. The mismatch can also occur due to direct current (DC) offset, local oscillator leakage, or other factors. As shown in FIG. 2, when there is a large mismatch between I+ and I−, a worst case IP2 is approximately 25 dBm is produced. The following example uses the worst case IP2 of 25 dBm from measured data of a known 3G receiver, such as the receiver 102.IMD2=Pin−(IP2−(Pin))=−25−(25−(−25))=−75 dBm=3.16 μwatts
When a transmit signal is at maximum power, 25 dBm, the IMD2 referred to the input of the transconductance amplifier (TCA) in FIG. 1, is −75 dBm. The received power spectral density, Ior, as per the 3rd Generation Partnership Project (3GPP) sensitivity specification, should be at or below −106.7 dBm to achieve a 0.1% bit error rate. The thermal noise, kTBF, in this example is approximately −99 dBm. Because the power of the IMD2 over the bandwidth of the desired signal is much greater than the kTBF, the sensitivity rises to −82.7 dB, i.e., 24 dB above the required sensitivity. The receiver 102, including, in particular, a direct-conversion receiver that lacks a SAW, should have a sufficiently large IP2 to meet the 3GPP specifications.
The IMD2 results from squaring the transmitted signal at the input to the mixer 114 and 115; therefore, without careful control of the IP2 of the mixer 114 and 115, the IMD2 could dominate the sensitivity of the receiver 102. FIG. 3 is a graph of the power of the Ior and the power of the IMD2, versus frequency, at the input of the A/D converter 130 in the receiver 102 when there is a large mismatch between I+ and I− that produces a relatively low IP2 of approximately 25 dBm. The top portion of FIG. 3 shows the power of the IMD2 from zero to 6 MHz. The bottom portion shows the power of the Ior. FIG. 3 graphically shows a relatively high IMD2 disadvantageously dominating the sensitivity of the receiver 102.
Most known methods for increasing the IP2 of a mixer in a direct-conversion receiver involve improving the selectivity of the analog line-up of the receiver. Other known methods for increasing the IP2 in a direct-conversion receiver use multiple receivers, or use a circuit that estimates the DC offset of the mixer, to improve the IP2.